Control device

ABSTRACT

In a sensorless control of a motor, due to the characteristic of a response frequency, an induced voltage, a magnetic pole position estimation gain, a current control gain, and a speed control gain are closely related. An object of the invention is to enable the induced voltage and the frequency characteristic of a magnetic pole position estimation system to be clearly designed and to theoretically and quantitatively design all of the control gains necessary for the sensorless control so as to solve a problem that a method of designing those parameters cannot be established and the parameters have to be adjusted in a try and error manner. A control device has an estimator estimating an estimation induced voltage and a phase error of a motor by applying an induced-voltage observer and a controller controlling the motor on the basis of the estimation induced voltage and the phase error.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a Continuation of U.S. patent application Ser. No.15/699,791 filed on Sep. 8, 2017, which claims the benefit of JapanesePatent Application No. 2016-177461 filed on Sep. 12, 2016 including thespecification, drawings and abstract are incorporated herein byreference in their entirety.

BACKGROUND

The present invention relates to a control device and relates to, forexample, a control device of a motor.

In a brushless DC motor, as one of general control methods, on the basisof a rotating coordinate system in which a rotor magnetic pole directionis defined as a d axis and an axis which is advanced by 90° in the θpositive direction from the d axis is defined as a q axis, a d-axiscurrent is set as 0, and linear control of a q-axis current and a torqueis performed.

A technique of computing the rotation position of a rotor in thebrushless DC motor on the basis of currents flowing in three-phasecoils, not detecting the rotation position by using a sensor, isproposed.

Since the magnetic pole position (the position of the dq axis) of therotor is unknown in a sensorless control including a method based on acurrent estimation error, the magnitude and the phase of an inducedvoltage generated in the motor are estimated and, from them, themagnetic pole position (the position of the dq axis) of the rotor isestimated.

For example, in Japanese Unexamined Patent Application Publication No.Hei 8(1996)-308286, current values Iu and Iv flowing in a stator coilare detected, and the rotation angle is estimated from the detectedvalues. In Non-Patent Literature 1, “Sensorless Brushless DC MotorControl Method Based on Current Estimation Error”, Takaharu Takeshita,Naofumi Nomura, and Nobuyuki Matsui, T. IEE Japan, Vol. 115-D, No. 4,1995, a sensorless brushless DC motor control based on a currentestimation error is described.

SUMMARY

Generally, in sensorless control of a motor based on a currentestimation error, the induced voltage and the frequency characteristicof the magnetic pole position estimation system depend on the speed ofthe motor. Consequently, when the rotation of the motor becomes lowspeed, a convergence gain of an estimation system becomes small, and theresponse frequency becomes lower.

In the sensorless control of the motor, to perform current control andcomputation of estimation of the induced voltage and the magnetic poleposition in a stable control axis, induced-voltage estimation responsetime has to be equal to or longer than current response time of acontrol system.

To maintain the relation, as the rotation of the motor decreases, theresponse frequency of current has to be decreased. Further, when theresponse frequency of current is decreased, in some cases, the responsefrequency of speed has to be decreased.

As described above, in the motor sensorless control, due to thecharacteristic of a response frequency, an induced voltage, a magneticpole position estimation gain, a current control gain, and a speedcontrol gain are closely related. However, there is a problem such thata method of designing those parameters cannot be established and theparameters have to be adjusted in a try and error manner.

The other problems and novel features will become apparent from thedescription of the specification and the appended drawings.

According to an embodiment, by using an induced-voltage observer, aninduced voltage is estimated, and all of control gains areunconditionally determined.

According to the embodiment, the induced voltage and the frequencycharacteristic of a magnetic pole position estimation system can beclearly designed, and all of control gains necessary for sensorlesscontrol can be designed theoretically and quantitatively.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram illustrating the configuration of a controldevice according to outline of an embodiment.

FIG. 2 is a block diagram illustrating the configuration of a controldevice according to a first embodiment.

FIG. 3 is a diagram for explaining the configuration of a motor and acoordinate system.

FIG. 4 is a diagram for explaining true coordinate axes desired to beestimated and coordinate axes used for control.

FIG. 5 is a block diagram of a d-axis induced-voltage estimation system.

FIG. 6 is a block diagram of a q-axis induced-voltage estimation system.

FIG. 7 is a diagram illustrating the details of an estimator in thecontrol device of the first embodiment.

FIG. 8 is a block diagram illustrating the configuration of a controldevice according to a second embodiment.

FIG. 9 is a block diagram of a voltage control system realizing deadtime compensation control.

FIG. 10 is a block diagram illustrating the configuration of a controldevice according to a third embodiment.

FIG. 11 is a graph illustrating an example of generation of a speedinstruction value from a position instruction value.

FIG. 12 is a graph illustrating an example of shift to a pull-inoperation mode.

FIG. 13 is a circuit diagram illustrating the configuration of aswitching unit in the control device of the third embodiment.

DETAILED DESCRIPTION

For clarification of description, omission and simplification areproperly made in the following description and drawings. Each ofelements illustrated in the drawings as function blocks performingvarious processes can be constructed by a CPU, a memory, and othercircuits as hardware, and is realized by a program loaded to a memory assoftware. Therefore, a person skilled in the art understands that thefunction blocks can be realized in various forms of only hardware, onlysoftware, or combination of the hardware and software, and the inventionis not limited to any of the forms. In the drawings, the same referencenumeral is designated to the same element and repetitive description isomitted as necessary.

Outline of Embodiments

FIG. 1 is a block diagram illustrating the configuration of a controldevice according to outline of an embodiment. In FIG. 1, a controldevice 100 has an estimator 101 and a controller 102. The control device100 controls a motor.

The estimator 101 estimates an estimated induction voltage and a phaseerror of a motor from a target voltage of the motor, a current value ofthe motor, and a magnetic pole position (rotation angle) of a rotor ofthe motor estimated in the past by applying an induced-voltage observer.

The controller 102 controls the motor on the basis of the estimatedinduction voltage and the phase error.

As described above, with the control device related to the outline ofthe embodiment, by estimating the induced voltage by using theinduced-voltage observer and unconditionally determining all of controlgains, the induced voltage and the frequency characteristic of amagnetic pole position estimation system can be clearly designed, andall of control gains necessary for sensorless control can be designedtheoretically and quantitatively.

First Embodiment

In a first embodiment, the detailed configuration of the control device100 described in the outline of embodiments and a control device of amotor using the control device 100 will be described.

First, the functions of components of the control device according tothe first embodiment will be described. FIG. 2 is a block diagramillustrating the configuration of a control device according to thefirst embodiment. In FIG. 2, a control device 200 has the estimator 101,the controller 102, a speed computing unit 201, a subtracter 202, a 2phase/3 phase converter 203, a PWM controller 204, and a 3 phase/2 phaseconverter 205. The control device 200 controls the rotation speed of amotor 210. For example, the estimator 101, the controller 102, the speedcomputing unit 201, and the subtracter 202 may be constructed by aprocessor or a semiconductor device made by an ASIC (ApplicationSpecific Integrated Circuit) or a CPU (Central Processing Unit) and amemory.

The estimator 101 estimates the present rotation angle of the rotor ofthe motor 210 on the basis of a U-phase current value Iu, a V-phasecurrent value Iv, and the rotation angle of the rotor of the motor 210in the past (for example, one sample before). The estimator 101 outputsthe estimated rotation angle to the speed computing unit 201, the 2phase/3 phase converter 203, and the 3 phase/2 phase converter 205. Thedetailed configuration and operation of the estimator 101 will bedescribed later.

The controller 102 computes target voltages V_(d) and V_(q) on the dqcoordinate axis on the basis of the deviation between a current targetvalue I_(q)* determined from the difference between a target rotationspeed and the rotation speed obtained by the speed computing unit 201and a 3-phase current actually flowing in the motor 210. The controller102 outputs the computation result to the estimator 101 and the 2phase/3 phase converter 203.

The speed computing unit 201 calculates the rotation speed on the basisof the rotation angle estimated by the estimator 101. The speedcomputing unit 201 outputs the obtained rotation speed to the subtracter202.

The subtracter 202 subtracts the rotation speed obtained by the speedcomputing unit 201 from the target rotation speed instructed from theoutside. The subtracter 202 outputs the subtraction result to thecontroller 102.

The 2 phase/3 phase converter 203 converts the target voltages V_(d) andV_(q) instructed from the controller 102 to target voltages Vu*, Vv*,and Vw* of actual three phases. The 2 phase/3 phase converter 203outputs the target voltages Vu*, Vv*, and Vw* of actual three phases tothe PWM controller 204.

The PWM controller 204 determines a duty ratio which realizes the targetvoltages Vu*, Vv*, and Vw* of three phases by on/off of anot-illustrated DC power supply. The PWM controller 204 drives the motor210 by a pulse wave of the duty ratio. For example, the PWM controller204 has therein an inverter, controls six switching elements, andcontrols voltages applied to coils of three phases in the motor 210. Theinverter has a current sensor detecting currents of the U phase and theV phase, and a U-phase current value Iu and a V-phase current value Ivare detected. The inverter may be provided on the inside or the outsideof the PWM controller 204.

The 3 phase/2 phase converter 205 calculates the current values I_(d)and I_(q) on the dq coordinate axis from the rotation angle estimated bythe estimator 101 and the current values Iu and Iv of the phasesdetected by the PWM controller 204. Then, the 3 phase/2 phase converter205 outputs the current values I_(d) and I_(q) to the estimator 101 andthe controller 102.

Next, the internal configuration of the estimator 101 will be described.In FIG. 2, the estimator 101 has an induced-voltage observer 111, aninduced-voltage computing unit 112, a phase computing unit 113, and asubtracter 114.

The induced-voltage observer 111 estimates magnitude “e” of anestimation induced voltage and a phase error β on the basis of thetarget voltages V_(d) and V_(q) on the dq coordinate axis, the currentvalues I_(d) and I_(q) on the dq coordinate axis, and the rotation angleof the rotor of the motor 210 estimated in the past. The induced-voltageobserver 111 outputs the magnitude “e” of the estimation induced voltageto the induced-voltage computing unit 112. The induced-voltage observer111 outputs the phase error β to the phase computing unit 113.

The induced-voltage computing unit 112 performs computation of dividingthe magnitude of the induced voltage by an induced-voltage coefficient.The induced-voltage computing unit 112 outputs the computation result tothe subtracter 114.

The phase computing unit 113 performs computation of integration ofmultiplying the phase error β by an error angle integration gain ωβ. Thephase computing unit 113 outputs the computation result to thesubtracter 114.

The subtracter 114 subtracts the computation result of the phasecomputing unit 113 from the computation result of the induced-voltagecomputing unit 112 to obtain a magnetic pole position estimation valueθ{circumflex over ( )}. The subtracter 114 outputs the magnetic poleposition estimation value θ{circumflex over ( )} to the induced-voltageobserver 111, the speed computing unit 201, the 2 phase/3 phaseconverter 203, and the 3 phase/2 phase converter 205.

As described above, in the estimator 101, the magnetic pole positionestimation value θ{circumflex over ( )} is controlled by adding phaseinformation calculated from the amplitude “e” and multiplying the phaseerror β with the error angle integration gain ωβ so that β becomes zero(β=0).

The internal configuration of the controller 102 will now be described.In FIG. 2, the controller 102 has a speed PI controller 121, subtracters122 and 123, a d-axis current PI controller 124, and a q-axis current PIcontroller 125.

The speed PI controller 121 determines a current target value I_(q)*determined from the difference between the target rotation speed and therotation speed obtained by the speed computing unit 201. The speed PIcontroller 121 outputs the current target value I_(q)* to the subtracter123.

The subtracter 122 subtracts the current value I_(d) output from the 3phase/2 phase converter 205 from the target value I_(d)* (=0). Thesubtraction result is output to the d-axis current PI controller 124.

The subtracter 123 subtracts the current value I_(q) output from the 3phase/2 phase converter 205 from the target value I_(q)*. Thesubtraction result is output to the q-axis current PI controller 125.

The d-axis current PI controller 124 determines a target voltage V_(d)from the difference between the target value I_(d)* and the currentvalue I_(d). The d-axis current PI controller 124 outputs the targetvoltage V_(d) to the 2 phase/3 phase converter 203.

The q-axis current PI controller 125 determines a target voltage V_(q)from the difference between the target value I_(q)* and the currentvalue I_(q). The q-axis current PI controller 125 outputs the targetvoltage V_(q) to the 2 phase/3 phase converter 203.

As described above, in the controller 102, on the basis of thedeviations between the current target value I_(q)* and the currentvalues I_(d) and I_(q) on the dq coordinate axis obtained by convertingthe currents of three phases actually flowing in the motor 210, thetarget voltages V_(d) and V_(q) on the dq coordinate axis are computed.

Next, the motor 210 will be described. FIG. 3 is a diagram forexplaining the configuration of the motor and a coordinate system. Inthis embodiment, the motor 210 is a three-phase brushless motor. Asillustrated in FIG. 3, the motor 210 has a rotor 220 as a field andstator windings 231, 232, and 233 of a U phase, a V phase, and a W phasedisposed in a stator 230 opposed to the rotor 220. The motor 210 may beof an inner rotor type in which a stator is disposed on the outside of arotor so as to face the rotor or an outer rotor type in which a statoris disposed on the inside of a cylindrical rotor so as to face therotor.

Three-phase fixed coordinates (UVW coordinate system) having a U axis, aV axis, and a W axis in the directions of the stator windings 231, 232,and 233 of the respective phases are defined. A two-phase rotationcoordinate system (the dq coordinate system or actual rotationcoordinate system) in which the d axis (magnetic pole axis) is set inthe magnetic pole direction of the rotor 220 and the q axis (torqueaxis) is set in a direction orthogonal to the d axis in the rotationplane of the rotor 220 is also defined. The dq coordinate system is arotation coordinate system which rotates together with the rotor 220. Inthe dq coordinate system, only the q-axis current contributes to torquegeneration of the rotor 220. Consequently, it is sufficient to set thed-axis current to zero and control the q-axis current in accordance witha desired torque. The rotation angle (rotor angle) θ of the rotor 220 isthe rotation angle of the d axis with respect to the U axis. The dqcoordinate system is an actual rotation coordinate system according tothe rotor angle θ. By using the rotor angle θ, the coordinate conversioncan be performed between the UVW coordinate system and the dq coordinatesystem.

In the first embodiment, the control device 100 controls the motor 210as the above-described three-phase brushless motor. Next, the operationof the control device 100 of the first embodiment will be described.

1. Estimation of Induced Voltage and Phase Using Induced-VoltageObserver

By using FIG. 4, estimation of the induced voltage and the phase usingthe induced-voltage observer will be described. FIG. 4 is a diagram forexplaining true coordinate axes desired to be estimated and coordinateaxes used for control. In FIG. 4, the d axis is a true d axis desired tobe estimated and the d{circumflex over ( )} axis is a d axis used forcontrol. Similarly, in FIG. 4, the q axis is a true q axis desired to beestimated, and the q{circumflex over ( )} axis is a q axis used forcontrol. In FIG. 4, θ indicates a true magnetic pole position, andθ{circumflex over ( )} denotes a magnetic pole estimation position. βdenotes an error angle between the d axis and the d{circumflex over ( )}axis, and an error angle between the q axis and the q{circumflex over( )} axis. “e” indicates an induced voltage appeared on the d{circumflexover ( )}q{circumflex over ( )} axis.

From FIG. 4, voltage equations on the dq control axis are written as thefollowing formulae (1) and (2).v* _(d)=(R+sL _(d))i _(d) −ω*L _(q) i _(q) +e _(d)  (1)v* _(q)=(R+sL _(q))i _(q) +ω*L _(d) i _(d) +e _(q)  (2)In the formulae (1) and (2), V_(d)* and V_(q)* are instruction voltagevalues of the d axis and the q axis. R denotes a winding resistancevalue of the coil of the rotor. “s” denotes a differential operator(Laplace operator). L_(d) and L_(d) are inductances in the d axis andthe q axis, respectively. i_(d) and i_(d) are current values in the daxis and the q axis, respectively. ω* denotes target rotation speed.e_(d) and e_(q) denote estimation induced voltage values in the d axisand the q axis, respectively.

Now, −ω*L_(q)i_(q) and −ω*L_(d)i_(d) are regarded as disturbances andset as −d_(d) and −d_(q).v* _(d)=(R+sL _(d))i _(d) −d _(d)  (3)v* _(q)=(R+sL _(q))i _(q) −d _(q)  (4)

Then, voltage equations become as the above formulae (3) and (4) andequations in which the d axis and the q axis are separated. First, anestimation formula of the d-axis induced voltage is derived. The formula(3) is rewritten to the following formula (5).

$\begin{matrix}{{si}_{d} = {\frac{v_{d}^{*}}{L_{d}} - {\frac{R}{L_{d}}i_{d}} + \frac{d_{d}}{L_{d}}}} & (5)\end{matrix}$

On the basis of the formula (5), i_(d) and d (disturbance) are set asstate variables, and a state equation (6) and a formula (7) are set.

$\begin{matrix}{{si}_{d} = {{{- \frac{R}{L_{d}}}i_{d}} + \frac{d}{L_{d}} + \frac{v_{d}^{*}}{L_{d}}}} & (6) \\{{sd} = {sd}_{d}} & (7)\end{matrix}$

When estimation values of i_(d) and d are set as

{circumflex over (d)}

and

, respectively, estimation state equations on the observer side can bewritten as the following formulae (8) and (9) by adding terms in whichestimation errors are multiplied by estimation gains K_(Ed1) andK_(Ed2).

$\begin{matrix}{{s} = {{- \frac{R}{L_{d}}} + \frac{\hat{d}}{L_{d}} + \frac{v_{d}^{*}}{L_{d}} + {K_{{Ed}\; 1}( {i_{d} -} )}}} & (8) \\{{s\;\hat{d}} = {K_{{Ed}\; 2}( {i_{d} -} )}} & (9)\end{matrix}$

When the formula (9) is substituted into the formula (8),

is expressed as the following formula (10).

$\begin{matrix}{= {\frac{\frac{K_{{Ed}\; 2}}{L_{d}}}{s^{2} + {( {\frac{R}{L_{d}} + K_{{Ed}\; 1}} )s} + \frac{K_{{Ed}\; 2}}{L_{d}}}\{ {{( {1 + {\frac{K_{{Ed}\; 1}}{K_{{Ed}\; 2}}L_{d}s}} )i_{d}} + {\frac{s}{K_{{Ed}\; 2}}v_{d}^{*}}} \}}} & (10)\end{matrix}$

When the formula (10) is substituted into the formula (9),

{circumflex over (d)}

can be written as the following formula (11).

$\begin{matrix}{\hat{d} = {= {\frac{\frac{K_{{Ed}\; 2}}{L_{d}}}{s^{2} + {( {\frac{R}{L_{d}} + K_{{Ed}\; 1}} )s} + \frac{K_{{Ed}\; 2}}{L_{d}}}\{ {{( {{L_{d}s} + R} )i_{d}} - v_{d}^{*}} \}}}} & (11)\end{matrix}$

FIG. 5 is a block diagram of a d-axis induced-voltage estimation systemto which the induced-voltage observer is applied. Concretely, FIG. 5 isa block diagram expressing the formula (11).

Looking at the formulae (10) and (11),

and become secondary systems of formulae (12) and (13) using i_(d) andv_(d)* as inputs.

$\begin{matrix}{\omega_{EG} = \sqrt{\frac{K_{{Ed}\; 2}}{L_{d}}}} & (12) \\{\zeta_{EG} = \frac{\frac{R}{L_{d}} + K_{{Ed}\; 1}}{\sqrt[2]{\frac{K_{{Ed}\; 2}}{L_{d}}}}} & (13)\end{matrix}$

That is, the frequency characteristics of the d-axis induced-voltageestimation system can be designed by a natural frequency ω_(EG) and adamping factor ζ_(EG), and estimation gains K_(Ed1) and K_(Ed2) can bewritten as the following formulae (14) and (15), respectively.

$\begin{matrix}{K_{{Ed}\; 1} = {{2\zeta_{EG}\omega_{EG}} - \frac{R}{L_{d}}}} & (14) \\{K_{{Ed}\; 2} = {\omega_{EG}^{2}L_{d}}} & (15)\end{matrix}$

Subsequently, similar calculation is performed also on the q axis. Theformula (4) is rewritten as the following formula (16).

$\begin{matrix}{{si}_{q} = {\frac{v_{q}^{*}}{L_{q}} - {\frac{R}{L_{q}}i_{q}} + \frac{d_{q}}{L_{q}}}} & (16)\end{matrix}$

On the basis of the formula (16), using i_(q) and d (disturbance) asstate variables, state equations (17) and (18) are set.

$\begin{matrix}{{si}_{q} = {{{- \frac{R}{L_{q}}}i_{q}} + \frac{d_{q}}{L_{q}} + \frac{v_{q}^{*}}{L_{q}}}} & (17) \\{{sd} = {sd}_{q}} & (18)\end{matrix}$

When estimation values of i_(q) and d are expressed as

{circumflex over (d)}

and, estimation state equations on the observer side can be written asthe following formulae (19) and (20) by adding terms in which anestimation error is multiplied by estimation gains K_(Eq1) and K_(Eq2).

$\begin{matrix}{{s} = {{- \frac{R}{L_{q}}} + \frac{\hat{d}}{L_{q}} + \frac{v_{q}^{*}}{L_{q}} + {K_{{Eq}\; 1}( {i_{q} -} )}}} & (19) \\{{s\hat{d}} = {K_{{Eq}\; 2}( {i_{q} -} )}} & (20)\end{matrix}$

When the formula (20) is substituted into the formula (19),

is expressed as the following formula (21).

$\begin{matrix}{= {\frac{\frac{K_{Eq2}}{L_{d}}}{s^{2} + {( {\frac{R}{L_{q}} + K_{{Eq}\; 1}} )s} + \frac{K_{Eq2}}{L_{q}}}\{ {{( {1 + {\frac{K_{{Eq}\; 1}}{K_{{Eq}\; 2}}L_{q}s}} )i_{q}} + {\frac{s}{K_{{Eq}\; 2}}v_{q}^{*}}} \}}} & (21)\end{matrix}$

When the formula (21) is substituted into the formula (20),

{circumflex over (d)}

can be written as the following formula (22).

$\begin{matrix}{\hat{d} = {= {\frac{\frac{K_{Eq2}}{L_{d}}}{s^{2} + {( {\frac{R}{L_{q}} + K_{{Eq}\; 1}} )s} + \frac{K_{Eq2}}{L_{q}}}\{ {{( {{L_{q}s} + R} )i_{q}} - v_{q}^{*}} \}}}} & (22)\end{matrix}$

FIG. 6 is a block diagram of a q-axis induced-voltage estimation systemto which the induced-voltage observer is applied. Concretely, FIG. 6 isa block diagram expressing the formula (11).

Looking now at the formulae (21) and (22),

and become secondary systems of formulae (23) and (24) using i_(q) andV_(q)* as inputs.

$\begin{matrix}{\omega_{EG} = \sqrt{\frac{K_{Eq2}}{L_{q}}}} & (23) \\{\zeta_{EG} = \frac{\frac{R}{L_{q}} + K_{{Eq}\; 1}}{\sqrt[2]{\frac{K_{E\;{q2}}}{L_{q}}}}} & (24)\end{matrix}$

That is, like in the d-axis induced-voltage estimation system, thefrequency characteristics of the q-axis induced-voltage estimationsystem can be also designed by ω_(EG) and ζ_(EG), and estimation gainsK_(Eq1) and K_(Eq2) can be written as the following formulae (25) and(26), respectively.

$\begin{matrix}{K_{{Eq}\; 1} = {{2\zeta_{EG}\omega_{EG}} - \frac{R}{L_{q}}}} & (25) \\{K_{{Eq}\; 2} = {\omega_{EG}^{2}L_{q}}} & (26)\end{matrix}$

Subsequently, when estimation induced-voltages are calculated from theestimated disturbances

and obtained by the formulae (11) and (22), the following formulae (27)and (28) are derived.e _(d) =−

+ω*L _(q) i _(q)  (27)e _(q) =−

−ω*L _(d) i _(d)  (28)

Therefore, the magnitude “e” of the estimation induced-voltage and thephase error β are obtained as the following formulae (29) and (30).

$\begin{matrix}{e = \sqrt{e_{d}^{2} + e_{q}^{2}}} & (29) \\{\beta = {{atan}( \frac{e_{d}}{e_{q}} )}} & (30)\end{matrix}$

Next, from the calculated “e” and β, the magnetic pole position isestimated. FIG. 7 is a diagram illustrating the details of the estimatorin the control device of the first embodiment. In FIG. 7, the functionof estimating the induced voltage and the magnetic pole position in thecontrol device of FIG. 2 is illustrated. In an induced-voltageestimation system 701, when an instruction voltage value and a detectioncurrent value are input, the phase error β and the magnitude “e” of theinduced voltage are output. A magnetic pole position estimation value OAis controlled by adding phase information calculated from the amplitude“e”, performing integration by multiplying the phase error β by an errorangle integration gain ω_(β) so that β becomes equal to 0 (β=0).

A frequency characteristic G_(E) of the induced-voltage estimationsystem becomes a stable secondary system determined by the naturalfrequency ω_(EG) and the damping factor ζ_(EG) when the estimation gainsK_(Ed1), K_(Ed2), K_(Eq1), and K_(Eq2) are designed as formulae (31),(32), (33), and (34), respectively.

$\begin{matrix}{K_{{Ed}\; 1} = {{2\zeta_{EG}\omega_{EG}} - \frac{R}{L_{d}}}} & (31) \\{K_{{Ed}\; 2} = {\omega_{EG}^{2}L_{d}}} & (32) \\{K_{{Eq}\; 1} = {{2\zeta_{EG}\omega_{EG}} - \frac{R}{L_{q}}}} & (33) \\{K_{{Eq}\; 2} = {\omega_{EG}^{2}L_{q}}} & (34)\end{matrix}$

Therefore, the control gains which have to be designed in the inducedvoltage and magnetic pole position estimation system are the threecontrol gains ω_(EG), ζ_(EG), and ω_(β).

2. Gain Designing Method

2-1. Design of Speed Control System and Current Control System

Bandwidth ω_(SC) of the speed control system and bandwidth ω_(CC) of thecurrent control system are designed to maintain the relation of theformula (35) so as not to interfere each other.ω_(CC)>>ω_(SC)  (35)2-2. Design of Induced-Voltage Estimation System

As illustrated in FIGS. 5 and 6, since the frequency characteristic ofthe induced-voltage estimation system is a secondary system determinedby ω_(EG) and ζ_(EG), ω_(EG) and ζ_(EG) are designed so that aninduced-voltage estimation response becomes equal to or higher than acurrent response.

According to the above-described design concept, ζ_(EG) is fixed to avalue of about 0.6 and ω_(EG) is designed so as to satisfy the relationof the following formula (36).ω_(EG)≥ω_(CC)  (36)2-3. Design of Magnetic Pole Position Estimation System

It is sufficient to design the error angle integration gain ω_(β) in themagnetic pole position estimation system in FIG. 7 to a valuesufficiently low relative to the bandwidth of the induced-voltageestimation system so that a phase margin can be sufficiently assured.However, when the error angle integration gain ω_(β) is set to be toolow, the bandwidth of the speed control system is high and, in the casewhere the magnetic pole position estimation system receives ahigh-frequency input, a large phase lag may occur.

Since the phase lag is directly related to torque decrease, the lowerlimit of the error angle integration gain ω_(β) is determined accordingto allowance of the phase lag.

According to the above-described design concept, the error angleintegration gain ω_(β) is designed. First, when the error angleintegration gain ω_(β) is designed as the following formula (37), thefrequency characteristic of the magnetic pole position estimation systembecomes almost the same as the characteristic of a first-order lagsystem, so that the phase lags about 45° at the cutoff frequency ω_(β).ω_(EG)>>ω_(β)  (37)

That is, it can be considered that the upper limit of the inputfrequency of the magnetic pole position estimation system is determinedby the bandwidth of the speed control system. Consequently, for example,when the bandwidth of the speed control system is designed to be equalto or less than the error angle integration gain ω_(β), an error isabout 45° at the maximum. Using 45° as the allowable range of an error(torque decrease rate is about 30%), the relation between the bandwidthof the speed control system and the error angle integration gain isdesigned so as to always maintain the relation of the following formula(38).ω_(β)>ω_(SC)  (38)

When the formulae (37) and (38) are combined, the following formula (39)is obtained.ω_(EG)>>ω_(β)≥ω_(SC)  (39)

When 2-1, 2-2, and 2-3 are combined, the relations of the gains of thecontrol systems are expressed as the formula (40).ω_(EG)≥ω_(CC)>>ω_(β)≥ω_(SC)  (40)

That is, when motor parameters are known, design of the entiresensorless control system using the induced-voltage observer is onlydesign of gains while maintaining the relations of the formula (40).

As described above, according to the control device of the firstembodiment, by applying the induced-voltage observer to theinduced-voltage estimation algorithm, the frequency characteristic ofthe induced-voltage estimation system can be clearly determined by thenatural frequency ω_(EG). The bandwidth ω_(CC) of the current controlsystem, the phase error integration gain ω_(β), and the bandwidth ω_(SC)of the speed control system are unconditionally determined according tothe formula (40), gain adjustment is unnecessary, and stable control canbe realized.

Second Embodiment

Generally, in sensorless control, at the time of low speed, low voltagecannot be correctly output due to voltage distortion caused by dead timenecessary for inverter control. As a result, induced-voltage and themagnetic pole position cannot be accurately estimated.

In a second embodiment, negative feedback control is performed so thatan output voltage of an inverter matches an instruction voltage withouta lag. By decreasing the voltage distortion caused by the dead time, theprecision of estimation of the induced voltage and the magnetic poleposition is increased.

FIG. 8 is a block diagram illustrating the configuration of a controldevice according to the second embodiment. In FIG. 8, the same numeralsare designated to the same components as those in FIG. 2 and theirdescription will not be repeated. In FIG. 8, a control device 800 has adead time compensator 801.

The 2 phase/3 phase converter 203 converts the target voltages V_(d) andV_(q) instructed from the controller 102 to target voltages Vu*, Vv*,and Vw* of actual three phases. The 2 phase/3 phase converter 203outputs the target voltages Vu*, Vv*, and Vw* of actual three phases tothe dead time compensator 801.

The dead time compensator 801 performs negative feedback control so thatthe target voltages Vu*, Vv*, and Vw* of three phases output from the 2phase/3 phase converter 203 match the instruction voltage without a lag.The detailed operation of the dead time compensator 801 will bedescribed later. For example, the dead time compensator 801 may beconstructed by an ASIC (Application Specific Integrated Circuit) or aprocessor or a semiconductor device made by a CPU (Central ProcessingUnit) and a memory.

The PWM controller 204 determines a duty ratio which realizes the targetvoltages of three phases output from the dead time compensator 801 byon/off of a not-illustrated DC power supply. The PWM controller 204drives the motor 210 by a pulse wave of the duty ratio. For example, thePWM controller 204 has therein an inverter, and controls six switchingelements to control voltages applied to three-phase coils in the motor210.

The detailed operation of the dead time compensator 801 will now bedescribed. FIG. 9 is a block diagram of a voltage control systemrealizing dead time compensation control. The input/outputcharacteristics of the voltage control system of FIG. 9 are expressed bythe following formula (41).

$\begin{matrix}{\frac{V_{o}}{V^{*}} = \frac{K_{P} + \frac{K_{I}}{s}}{1 + \frac{K_{P} + \frac{K_{I}}{s}}{1 + \frac{s}{\omega_{C}}}}} & (41)\end{matrix}$

In the formula (41), K_(P) and K_(I) are a proportional gain and anintegration gain, respectively, in PI control. ω_(C) denotes cutofffrequency of a filter cutting frequencies below the frequency of a PWM.

When it is designed that K_(P)=1 and K_(I)=ω_(C), the formula (41)becomes the following formula (42).

$\begin{matrix}{\frac{V_{o}}{V^{*}} = {\frac{1 + \frac{\omega_{C}}{s}}{1 + \frac{1 + \frac{\omega_{C}}{s}}{1 + \frac{s}{\omega_{C}}}} = 1}} & (42)\end{matrix}$

That is, as expressed by the relation of the formula (42), aninstruction voltage V* and an output voltage V_(O) match without a lag.The dead time compensator 801 performs the control process of theformula (42).

As described above, according to the control device of the secondembodiment, the voltage distortion caused by dead time can be decreased,so that the precision of estimating the induced voltage and the magneticpole position increases, and the control can be performed at lowerspeed. In addition, since the voltage distortion caused by dead time canbe decreased, the lower limit speed of magnetic pole position estimationcan be lowered.

Third Embodiment

In a third embodiment, when the speed is below the lower limit speed ofestimation of the induced voltage and the magnetic pole position, themode is switched to a mode of performing a drive by passing d-axiscurrent and forcedly changing the phase in accordance with a positioninstruction value (this mode will be called a pull-in operation mode).

FIG. 10 is a block diagram illustrating the configuration of a controldevice according to the third embodiment. In FIG. 10, the same numeralsare designated to the same components in FIGS. 2 and 8 and theirdescription will not be repeated. In FIG. 10, a control device 1000 hasa switching unit 1001.

When rotation speed estimated by the speed computing unit 201 is largerthan a predetermined threshold, the switching unit 1001 outputs i_(d)*=0to the d-axis current PI controller 124 and outputs i_(q)* output fromthe speed PI controller 121 to the q-axis current PI controller 125.When the rotation speed is equal to or less than the predeterminedthreshold, the switching unit 1001 outputs i_(d)*=i_(d_ol) to the d-axiscurrent PI controller 124 and outputs i_(q)*=0 to the q-axis current PIcontroller 125. For example, the switching unit 1001 may be constructedby an ASIC (Application Specific Integrated Circuit) or a processor or asemiconductor device made by a CPU (Central Processing Unit) and amemory.

The operation of the control device 1000 will now be described. First,an operation of generating a speed instruction value from a positioninstruction value will be described. FIG. 11 is a graph illustrating anexample of generation of a speed instruction value from a positioninstruction value. In FIG. 11, the horizontal axis indicates time, andthe vertical axis indicates rotation angle (rotation position) androtation speed. In FIG. 11, θ* indicates target position, Ta indicatesacceleration time, ω_(MAX) indicates maximum speed, and ω* indicates aspeed instruction value.

The speed instruction value is generated by differential of the positioninstruction value. The speed instruction value depends on an inputparameter and becomes a trapezoid or triangle pattern. According to thespeed instruction value, the motor is driven.

Next, the operation of the control device 1000 in the case where themotor rotates at low speed will be described.

FIG. 12 is a graph illustrating an example of shift to a pull-inoperation mode. In FIG. 12, the horizontal axis indicates time, and thevertical axis indicates rotation speed, q-axis current, and d-axiscurrent. In FIG. 12, ω_(switch) denotes lower limit speed (speed of thethreshold of shift to the pull-in operation mode). i_(d_ol) indicatesd-axis current at the time of the pull-in operation. T_(switch)indicates shift time to the pull-in operation.

As illustrated in FIG. 12, when the rotation speed is higher than thelower limit speed ω_(switch), i_(q)* output from the speed PI controller121 is output to the q-axis current PI controller 125, and i_(d)*=0 isoutput to the d-axis current PI controller 124 to control the motor.When the rotation speed becomes equal to or less than the lower limitspeed ω_(switch), the d-axis current i_(d_ol) when i_(d)* is in thepull-in operation is output to the d-axis current PI controller 124within the shift time T_(switch) to the pull-in operation, and i_(q)*=0is output to the d-axis current PI controller 124 to control the motor.The d-axis current i_(d_ol) at the time of the pull-in operation ispreferably set to q-axis current just before switching of the mode.

By the above operation, at the time of switch to the pull-in operationmode, the q-axis current just before switching of the mode is passed tothe d-axis current PI controller 124, and the d-axis current and theq-axis current are replaced, thereby coupling the torques seamlessly.

Next, the internal configuration of the switching unit 1001 will bedescribed. FIG. 13 is a circuit diagram illustrating the configurationof the switching unit in the control device of the third embodiment. InFIG. 13, the switching unit 1001 has a determining unit 1011, a firstswitch 1021, and a second switch 1031.

The determining unit 1011 determines whether the rotation speedestimated by the speed computing unit 201 is larger than a predeterminedthreshold or is equal to or less than the predetermined threshold andoutputs the determination result to the first switch 1021 and the secondswitch 1031.

According to the determination of the determining unit 1011, the firstswitch 1021 couples any of contact points 1022 and 1023 to a contactpoint 124. To the contact point 1024, the d-axis current PI controller124 is coupled.

Concretely, when the rotation speed estimated by the speed computingunit 201 is larger than the predetermined threshold, the contact point1022 is coupled to the contact point 1024. That is, i_(d)*=0 is outputto the d-axis current PI controller 124.

When the rotation speed estimated by the speed computing unit 201 isequal to or less than the predetermined threshold, the contact point1023 is coupled to the contact point 1024. That is, i_(d)*=i_(d_ol) isoutput to the d-axis current PI controller 124. i_(d_ol) is the d-axiscurrent at the time of the pull-in operation.

According to the determination of the determining unit 1011, the secondswitch 1031 couples any of the contact points 1022 and 1033 to a contactpoint 1034. To the contact point 1032, the speed PI controller 121 iscoupled. To the contact point 1034, the q-axis current PI controller 124is coupled.

Concretely, when the rotation speed estimated by the speed computingunit 201 is larger than the predetermined threshold, the contact point1032 is coupled to the contact point 1034. That is, i_(q)* output fromthe speed PI controller 121 is output to the q-axis current PIcontroller 125.

When the rotation speed estimated by the speed computing unit 201 isequal to or less than the predetermined threshold, the contact point1033 is coupled to the contact point 1034. That is, i_(q)*=0 is outputto the q-axis current PI controller 125.

With the above configuration, in combination with the dead timecompensating function realized by negative-feedback-controlling thevoltage applied to the motor, drive is performed so as to follow theinstruction speed generated from the position instruction profile.

As described above, according to the control device of the thirdembodiment, by seamlessly switching the speed control mode and thepull-in operation mode using the lower limit speed as a threshold, andperforming driving on the basis of the instruction speed generated fromthe position instruction value, the simple positioning operation can berealized.

In the case where the state shifts from the state in which the speedexceeds the lower limit speed of estimation of the induced voltage andthe magnetic pole position to the state in which the speed is below thelower limit speed of estimation of the induced voltage and the magneticpole position, by passing the q-axis current just before the stateshifts to the d-axis current PI controller 124 and replacing the d-axiscurrent and the q-axis current, the torques can be seamlessly coupledand, in a state where the speed is below the lower limit speed, thepositioning operation can be realized.

Concretely, when the speed is below the lower limit speed, by changingthe mode to the pull-in operation mode and forcedly turning the phase inaccordance with the position instruction profile, the simple positioningoperation can be realized.

The above-descried program is stored by using any of non-transitorycomputer readable media of various types and can be supplied to acomputer. The non-transitory computer readable media include tangiblestorage media of various types. Examples of the non-transitory computerreadable media include magnetic recording media (for example, flexibledisk, magnetic tape, and hard disk drive), magnet-optic recording media(for example, magnet-optic disk), CD-ROM (Read Only Memory), CD-R,CD-R/W, and semiconductor memories (for example, mask ROM, PROM(Programmable ROM), EPROM (Erasable PROM), flash ROM, and RAM (RandomAccess Memory)). The program may be supplied to a computer by any oftransitory computer readable media of various types. Examples of thetransitory computer readable media include an electric signal, anoptical signal, and electromagnetic wave. The transitory computerreadable medium can supply a program to a computer via a wiredcommunication path such as an electric wire or an optical fiber or awireless communication path.

Although the present invention achieved by the inventors herein has beenconcretely described above on the basis of the embodiments, obviously,the present invention is not limited to the forgoing embodiments but canbe variously changed without departing from the gist.

For example, although the example of controlling the three-phasebrushless motor has been described in the foregoing embodiments, theinvention can be also applied to a stepping motor of a PM (PermanentMagnet) type or an HB (Hybrid) type using a permanent magnet other thana three-phase motor.

What is claimed is:
 1. A control device, comprising: an estimatorestimating an estimation induced voltage and a phase error of a motor byapplying an induced-voltage observe circuit, and a controllercontrolling the motor on the basis of the estimation induced voltage andthe phase error, wherein the estimator comprises the induced-voltageobserve circuit, an induced-voltage computing circuit and a phasecomputing circuit, wherein the induced-voltage computing circuitdividing the estimation induced voltage by an induced-voltagecoefficient, wherein the phase computing circuit performs integration bymultiplying the phase error with an error angle integration gain,wherein the estimator comprises a first subtractor, wherein the firstsubtractor obtains a magnetic pole position estimation value bysubtracting a computation result of the phase computing circuit from acomputation result of the induced-voltage computing circuit.
 2. Thecontrol device according to claim 1, wherein the induced-voltage observecircuit estimates the estimation induced voltage and the phase error ofthe motor on the basis of the magnetic pole position estimation value ofthe motor fed back from the first subtractor, a target voltage, and aresponse current.
 3. The control device according to claim 2, furthercomprising: a speed computing circuit calculating a rotation speed fromthe magnetic pole position estimation value; and a second subtractorsubtracting the rotation speed calculated by the speed computing circuitfrom a target rotation speed, wherein the controller controls the motoron the basis of a subtraction result of the second subtractor.
 4. Thecontrol device according to claim 2, further comprising: a PWMcontroller having an inverter and driving the motor by a pulse wave of aduty ratio on the basis of the target voltage, and a dead timecompensator performing a negative feedback control on the PWM controlleron the target voltage so that an output voltage of the inverter matchesan instruction voltage without a lag.
 5. The control device according toclaim 3, wherein the controller comprises: a speed PI controllerdetermining a current target value determined from a difference betweenthe target rotation speed and the rotation speed calculated by the speedcomputing circuit; a switching unit that: when the rotation speedcalculated by the speed computing circuit is larger than a predeterminedthreshold, sets a d-axis target current to zero and setting a q-axistarget current to the current target value determined from a differencebetween the target rotation speed and the rotation speed obtained by thespeed computing circuit; and when the rotation speed calculated by thespeed computing circuit is equal to or less than the predeterminedthreshold, sets a predetermined d-axis current at a time of a pull-inoperation and setting a q-axis target current to zero; a d-axis currentPI controller determining a d-axis target voltage from a differencebetween a d-axis target value and a d-axis current value; and a q-axiscurrent PI controller determining a q-axis target voltage from adifference between a q-axis target value and a q-axis current value. 6.The control device according to claim 3, wherein a predetermined d-axiscurrent at the time of the pull-in operation is a q-axis current valuewhen the rotation speed calculated by the speed computing circuitbecomes equal to or less than a predetermined threshold.